Method for starting a discharge lamp using high energy initial pulse

ABSTRACT

The described DC to AC inverter efficiently controls the amount of electrical power used to drive a cold cathode fluorescent lamp (CCFL). Additionally, during striking of the CCFL, a higher energy initial energy pulse is used. During normal operation, a lower energy pulse is used.

RELATED APPLICATIONS

This is a continuation-in-part of pending U.S. patent application Ser.No. 09/885,244 filed Jun. 19, 2001, which is a division U.S. patentapplication Ser. No. 09/528,407, now U.S. Pat. No. 6,316,881, which is adivision of U.S. patent application Ser. No. 09/209,586, now U.S. Pat.No. 6,114,814, priority from which is claimed under 35 U.S.C. §120.

FIELD OF THE INVENTION

The present invention relates to the field of discharge lighting and, inparticular, to efficiently supplying electrical power for driving adischarge lamp by initially starting the discharge lamp with a highenergy initial pulse.

BACKGROUND OF THE INVENTION

A discharge lamp, such as a cold cathode fluorescent lamp (CCFL), hasterminal voltage characteristics that vary depending upon the immediatehistory and the frequency of a stimulus (AC signal) applied to the lamp.Until the CCFL is “struck” or ignited, the lamp will not conduct acurrent with an applied terminal voltage that is less than the strikevoltage. Once an electrical arc is struck inside the CCFL, the terminalvoltage may fall to a run voltage that is approximately ⅓ of the strikevoltage over a relatively wide range of input currents. When the CCFL isdriven by an AC signal at a relatively high frequency, the CCFL (oncestruck) will not extinguish on each cycle and will exhibit a positiveresistance terminal characteristic. Since the CCFL efficiency improvesat relatively higher frequencies, the CCFL is usually driven by ACsignals having frequencies that range from 50 KiloHertz to 100KiloHertz.

Driving a CCFL with a relatively high frequency square-shaped AC signalwill produce the maximum useful lifetime for the lamp. However, sincethe square shape of an AC signal may cause significant interference withother circuits in the vicinity of the circuitry driving the CCFL, thelamp is typically driven with an AC signal that has a less than optimalshape such as a sine-shaped AC signal.

Most small CCFLs are used in battery powered systems, e.g., notebookcomputers and personal digital assistants. The system battery supplies adirect current (DC) voltage ranging from 7 to 20 Volts with a nominalvalue of about 12V to an input of a DC to AC inverter. A commontechnique for converting a relatively low DC input voltage to a higherAC output voltage is to chop up the DC input signal with power switches,filter out the harmonic signals produced by the chopping, and output arelatively clean sine-shaped AC signal. The voltage of the AC signal isstepped up with a transformer to a relatively high voltage, e.g., from12 to 1500 Volts. The power switches may be bipolar junction transistors(BJT) or field effect transistors (MOSFET). Also, the transistors may bediscrete or integrated into the same package as the control circuitryfor the DC to AC converter.

Since resistive components tend to dissipate power and reduce theoverall efficiency of a circuit, a typical harmonic filter for a DC toAC converter employs inductive and capacitive components that areselected to minimize power loss, i.e., each of the selected componentsshould have a high Q value. A second-order resonant filter formed withinductive and capacitive components is also referred to as a “tank”circuit because the tank stores energy at a particular frequency.

The electronic device incorporating the CCFL often is used in a widevariety of environmental conditions, such as wide temperaturevariations. Further, variations in the values of the components bothwithin the driving circuit and in external components normally occur.Because of this, the amount of energy needed to most efficiently strikethe CCFL may vary.

BRIEF DESCRIPTION OF THE DRAWINGS

The foregoing aspects and many of the attendant advantages of thisinvention will become more readily appreciated as the same becomesbetter understood by reference to the following detailed description,when taken in conjunction with the accompanying drawings, wherein:

FIG. 1A is an exemplary schematic of a power controlled integratedcircuit coupled to a tank circuit on a primary side of a step-uptransformer for driving the discharge lamp;

FIG. 1B is an exemplary schematic of a current controlled integratedcircuit coupled to another tank circuit on a primary side of the step-uptransformer for driving the discharge lamp;

FIG. 2A is an exemplary schematic of the power controlled integratedcircuit using a tank circuit disposed on the primary side of the step-uptransformer to drive the discharge lamp;

FIG. 2B is another exemplary schematic of the power controlledintegrated circuit using another tank circuit disposed on the secondaryside of the step-up transformer used to drive the discharge lamp;

FIG. 2C is another exemplary schematic of the power controlledintegrated circuit using another tank circuit disposed on the secondaryside of the step-up transformer employed to drive the discharge lamp;

FIG. 2D is another exemplary schematic of another tank circuit disposedon the secondary side of the step-up transformer used to drive thedischarge lamp;

FIG. 2E is another exemplary schematic of another tank circuit thatemploys a primary coupling capacitor;

FIG. 3A is an exemplary schematic of a power control integrated circuitfor driving the discharge lamp;

FIG. 3B is an exemplary schematic of a current controlled integratedcircuit for driving the discharge lamp;

FIG. 4 is an exemplary schematic of a power control block implemented bythe power control integrated circuit;

FIG. 5 is an exemplary schematic of a gate drive block implemented bythe current control and power control integrated circuits;

FIG. 6 is an exemplary overview of the various phases of the oscillationcycle of the invention;

FIGS. 7A-7D displays four graphs for the corresponding voltage andcurrent waveforms that are generated when driving the discharge lamp atboth maximum and partial duty cycle,

FIGS. 7E-7F illustrates two graphs for leading edge modulation of thevoltage waveform and the corresponding current waveform at partialpower;

FIGS. 8A and 8B shows two graphs for double sided modulation of thevoltage waveform and the corresponding current waveform at partialpower;

FIGS. 9A-9D illustrates four graphs for pulse train phase modulation ofthe voltage waveform and the current waveform at full power;

FIGS. 9E-9H displays four graphs for pulse train phase modulation of thevoltage waveform and the current waveform at partial power;

FIG. 10 shows the four states of the power switches and the direction ofthe load current during phase modulation; and

FIG. 11 is a flow diagram illustrating the method of the presentinvention.

DETAILED DESCRIPTION OF THE INVENTION

As noted above, inverters for driving a CCFL typically comprise a DC toAC converter, a filter circuit, and a transformer. Examples of suchcircuits are shown in U.S. Pat. No. 6,114,614 to Shannon et al.,assigned to the assignee of the present invention and hereinincorporated by reference in its entirety. In addition, other prior artinverter circuits, such as a constant frequency half-bridge (CFHB)circuit or a inductive-mode half-bridge (IMB) circuit, may be used todrive a CCFL. The present invention may be used in conjunction with anyof these inverter circuits, as well as other inverter circuits.

The disclosure herein teaches a method and apparatus for striking andsupplying electrical power to a discharge lamp, such as a cold cathodefluorescent lamp (CCFL). In accordance with the invention, the initialpulse of energy provided by the apparatus to the CCFL is larger than thesteady state energy pulses provided to the CCFL after the CCFL has beenstruck. In one embodiment, the initial pulse is made larger by wideningthe time of the pulse. In another embodiment, the initial pulse may bemade larger by increasing the voltage of the pulse, while maintainingthe width of the pulse. The important consideration is that the initialpulse has a higher energy content. It should further be noted that themethod of the present invention is described in connection with one typeof inverter. However, the method may be used with other inverters.

In one embodiment, the present invention is an integrated circuit (IC)that includes four power MOSFETs arranged in an H-bridge circuit. The ICin combination with a separate output network inverts a direct current(DC) signal into an alternating current (AC) signal with enough voltageto drive a load such as a discharge lamp. The IC drives the load at theresonant frequency of the output network in combination with the load'scapacitive and inductive components for a range of voltages that areprovided by a DC power source.

The H-bridge circuit generates an AC signal by periodically inverting aDC signal. The control circuitry regulates the amount of electricalpower delivered to the load by modulating the pulse width (PWM) of eachhalf cycle of the AC signal. Since the PWM provides for a symmetrical ACsignal during normal operation, even harmonic frequencies in the ACsignal are canceled out. By eliminating the even harmonics and generallyoperating at the resonant frequency of the filter (load), the designedloaded Q value of the filter may be fairly low and losses in the filtermay be minimized. Also, since the CCFL is connected directly across thesecondary winding of the step-up transformer, except for the fraction ofa second required to strike an arc inside the lamp, the step-uptransformer's secondary winding generally operates at the run voltage ofthe CCFL. Further, it will be seen further below that the controlcircuitry will selectively increase the width of the pulses provided tothe load during striking of the load, relative to normal operation.

Turning now to FIG. 1A, an exemplary schematic 100 displays the powercontrol embodiment of an integrated circuit 104 (IC) coupled to a loadthat includes a tank circuit 108 and a lamp 106 such as a CCFL. A DCpower supply 102, i.e., a battery, is connected to IC 104. A boostcapacitor 120 a is connected between a B STR terminal and an outputterminal 110 a, which is connected to another terminal labeled as OUTR.Similarly, another boost capacitor 120 b is connected between a BSTLterminal and an output terminal 110 b that is connected to anotherterminal identified as OUTL. The boost capacitors 120 a and 120 b areenergy reservoirs that provide a source of power to operate circuitryinside the IC 104 that can float above the operating voltage of the restof the circuitry.

An end of inductor 116 is connected to the output terminal 110 a and anopposite end of the inductor is coupled to an end of a capacitor 118 andan end of a primary winding of a step-up transformer 114. An oppositeend of the capacitor 118 is coupled to another end of the primarywinding of the step-up transformer 114 and the output terminal 110 b. Anend of a secondary winding for the step-up transformer 114 is connectedto a lamp terminal 112 a and another end of the secondary winding isconnected to a lamp terminal 112 b.

A reactive output network or the “tank” circuit 108 is formed by thecomponents connected between the output terminals 110 a and 110 b andthe primary winding of the step-up transformer 114. The tank circuit isa second-order resonant filter that stores electrical energy at aparticular frequency and discharges this energy as necessary to smooththe sinusoidal shape of the AC signal delivered to the lamp 106. Thetank circuit is also referred to as a self-oscillating circuit.

In FIG. 1B, an exemplary schematic 100′ displays the current controlembodiment of an IC 104′ coupled to a load that includes the tankcircuit 108 and the lamp 106. The schematic 100′ is substantiallysimilar to that of FIG. 1A, except that additional current sensing isincluded. Note that the second terminal of the secondary winding isdirectly connected to ground. The other lamp terminal 112 b is coupledto an anode of a diode 107 and a cathode of a diode 105. The cathode ofthe diode 107 is coupled to an end of a sense resistor 109 and a Vsenseterminal at the IC 104′. The anode of the diode 105 is coupled to theother end the sense resistor 109 and ground. In this case, the IC 104′monitors the voltage across the sense resistor 109 so that the amount ofcurrent flowing into the lamp 106 may be approximated and used tocontrol the amount of electrical power used to drive the lamp.

Additionally, it is envisioned that the power and current controlembodiments of the invention, i.e., IC 104 and IC 104′, may be used witha plurality of different embodiments of the tank circuit. In FIG. 2A,the tank circuit 108 shown in FIG. 1A and 1B is shown coupled to the IC104. The tank circuit 108 operates as a filter which is coupled to theprimary winding of the step-up transformer 114.

In FIG. 2B, another embodiment of a tank circuit 108′ is shown. One endof the primary winding for the step-up transformer 114 is connected tothe output terminal 110 a and the other end of the primary winding isconnected to the other output terminal 110 b. An end of an inductor 116′is coupled to one end of the secondary winding for the step-uptransformer and another end of the inductor is connected to an end ofcapacitor 118′ and the lamp terminal 112 a. The other end of thesecondary winding for the step-up transformer is coupled to another endof the capacitor 118′ and the other lamp terminal 112 b. In thisembodiment, the tank circuit 108′ has all of the reactive componentsthat form the second order filter disposed on the secondary winding sideof the step-up transformer 114.

FIG. 2C shows another embodiment of the tank circuit 108″ that issimilar to the tank circuit 108′ illustrated in FIG. 2B. However, thetank circuit 108″ does not employ a discrete inductive component to formthe second order filter for the tank. Instead, this embodiment uses aninherent leakage inductance 117 of the windings in the step-uptransformer 114 as the inductive element of the second order filter. Theelimination of a discrete inductive component to implement the secondorder filter of the tank circuit 108″ reduces cost.

FIG. 2D illustrates yet another embodiment of the tank circuit 108′″ forfurther reducing the cost to implement the present invention. In thisembodiment, the tank circuit 108′″ uses a parasitic capacitance 122 ofthe lamp wiring (largest source), * the secondary winding of the step-uptransformer 114 and the transformer's inherent inductance 117 to formthe second order filter. One end of the secondary winding fortransformer 114 is connected to the lamp terminal 112 a and the otherend of the secondary winding is connected to the lamp terminal 112 b.This embodiment eliminates the need for discrete inductive andcapacitive components to implement a second order filter.

FIG. 2E shows another embodiment of the tank circuit 108′″ that issubstantially similar to the embodiment shown in FIG. 2D. However, inthis case, the primary of the transformer 114 is coupled to the outputof the IC 104 through a capacitor 111 which is used to cancel out theeffect of the transformer's primary magnetizing inductance. The additionof the capacitor 111 causes the resonant frequency at the primarywindings of the transformer 114 to more closely match the resonantfrequency at the secondary winding of the transformer. In this way, theresonant frequency for the entire circuit, i.e., the tank circuit 108′″and the transformer 114, is brought closer to the resonant frequency atthe secondary windings of the transformer.

Additionally, the largest source of parasitic capacitance for thevarious tank circuits shown in FIGS. 2A-2E is the wiring for thedischarge lamp 106. It is also envisioned that a pair of parallel metalplates may be disposed on either side of a circuit board that includesthe IC 104 so that a capacitive component is formed for the second orderfilter (tank circuit).

FIGS. 3A, 3B, 4 and 5 illustrate the internal circuitry of an integratedcircuit (IC) for implementing the different embodiments of theinvention. FIG. 3A shows an exemplary schematic of the power controlembodiment of the IC 104. A Vref signal is provided as an output from avoltage regulator 124 a that is coupled to a Vsupply signal. The Vrefsignal is a bandgap reference voltage which is nominally set to 5.0Volts and it is used to derive various voltages used by separatecomponents of the IC 104. Several internal voltages for a control logicblock 146 are derived from the Vref signal, such as an UVLO(undervoltage lockout) signal and a master voltage reference for athermal shutdown circuit. Also, the Vref signal derives other voltagesthat set trip points for a peak current (Ipk) comparator 138, a zerocrossing detector 140 and a power control block 136.

A voltage regulator 124 b is also coupled to the Vsupply signal andprovides a regulated 6 Volt DC signal. The output of voltage regulator124 b is connected to a gate drive block 128 b and an anode of a diode126 a whose cathode is connected to a gate drive block 128 a and theBOOST LEFT terminal. Another voltage regulator 124 c is coupled to theVsupply signal and it provides a regulated 6 Volt DC signal to a gatedrive block 128 d. The output of the voltage regulator 124 c is alsoconnected to an anode of a diode 126 b whose cathode is connected to agate drive block 128 c and the BOOST RIGHT terminal. Since the voltageregulators 124 b and 124 c separately regulate the voltage supplied tothe relatively high power gate drive blocks 128 a, 128 b, 128 c and 128d, the operation of any of the gate drive blocks tends to notsignificantly interfere with the Vref signal outputted by the voltageregulator 124 a. Also, separate terminals for the gate drive blocks 128b and 128 d are connected to earth ground.

Two level shift amplifiers 132 a and 132 b, have their respective inputterminals separately connected to a control logic block 146 and theirouput terminals separately coupled to the gate drive blocks 128 a and128 c, respectively. These level shift amplifiers translate the controllogic signals from the logic level used in the control logic block 146to the logic levels required by the gate drive blocks 128 a and 128 c,respectively.

An H-bridge output circuit for IC 104 is defined by the four powerMOSFETs 130 a, 130 b, 130 c and 130 d. The drain terminal of the MOSFET130 a is coupled to the Vsupply signal and its gate terminal is coupledto gate drive block 128 a. The source terminal of the MOSFET 130 a isconnected to the OUT LEFT terminal, the gate drive block 128 a, thedrain terminal of the MOSFET 130 b, the gate drive block 128 b, and amux block 134. The source terminal of the MOSFET 130 b is connected toearth ground and its gate terminal is coupled to the gate drive block128 b. Similarly, the drain terminal of the MOSFET 130 c is connected tothe Vsupply signal and its gate terminal is coupled to the gate driveblock 128 c. The source terminal of the MOSFET 130 c is connected to theOUT RIGHT terminal, the gate drive block 128 c, the drain terminal ofthe MOSFET 130 d, the gate drive block 128 d, and the mux block 134.Also, the source terminal of the MOSFET 130 d is connected to ground andits gate terminal is coupled to the gate drive block 128 d.

The source terminals of the MOSFETs 130 b and 130 d are coupled to earthground (low side) and their respective gate drive blocks 128 b and 128 dinclude discrete digital logic components that employ a 0 to 5 Voltsignal to control the operation of the associated power MOSFETs. Thesource terminals of the MOSFETs 130 a and 130 c are not connected toearth ground. Instead, these source terminals are connected to therespective OUT LEFT and OUT RIGHT terminals (high side) of the H-Bridgeoutput circuit. In this arrangement, a 0 (ground) to 5 Volt signal maynot reliably control the operation of the MOSFETs 130 a and 130 c. Sincethe gate drive blocks 128 a and 128 c employ discrete digital logiccontrol signals, the invention provides for level shifting these controlsignals to a voltage that is always higher than the voltage at thesource terminals of the associated MOSFETs 130 a and 130 c. The sourceterminal voltages tend to rise along with the voltage impressed acrossthe OUT LEFT and OUT RIGHT terminals of the H-bridge output circuit. Thelevel shift amplifiers 132 a and 132 b translate a 0 to 5 Volt logicsignal that is referenced to ground into a 0 to 6 Volt logic signalwhich is referenced to the source terminal of the associated MOSFETs 130a and 130 c. In this way, when the source terminals of the MOSFETs 130 aand 130 c have a potential anywhere between 0 Volts and 25 Volts, thegate drive blocks 128 a and 128 c are still able to control theoperation of their associated MOSFETs.

The gate drive blocks 128 a, 128 b, 128 c and 128 d along with the levelshift amplifiers 132 a and 132 b translate the control signals from thecontrol block 146 into a drive signal for each of their associated powerMOSFETs in the H-bridge output circuit. The gate drive blocks providebuffering (current amplification), fault protection, level shifting forthe power MOSFET control signals, and cross conduction lockout. The gatedrive blocks amplify the current of the digital logic signals so thatrelatively high currents may be provided for the rapid switching of thestate of the power MOSFETs between the on (conduction) and off(non-conduction) states. Each of the four power MOSFETs is currentlimited by its associated gate drive blocks to approximately 5 Ampereswhen an output fault occurs such as a short from the OUT LEFT terminaland/or the OUT RIGHT terminal to the Vsupply rail or a short to earthground. Under such an output fault condition, the gate drive block willturn off the associated power MOSFET before any damage occurs.

All four of the power transistors in the preferred embodiment areMOSFETs, and they tend to have a high input capacitance. To quicklyswitch a power MOSFET between the conduction and non-conduction states,the gate drive block provides for driving large currents into the gateterminal of the respective power MOSFET. The gate drive blocks amplifythe small currents available from control signals produced by thediscrete digital logic elements in the blocks to a relatively highercurrent level that is required to quickly switch the state of the powerMOSFETs.

When the gate drive block applies a voltage signal (6 volts with respectto the source terminal) to the associated power MOSFET's gate terminal,the MOSFET will turn on (conduct). Also, the power MOSFET will turn off(non-conduct) when zero volts is applied to its gate terminal withrespect to its source terminal. In contrast, the source terminals of thetwo power MOSFETs 130 a and 130 c are connected to the respective leftoutput and right output terminals. This configuration causes the sourceterminal voltage to float for each of these power MOSFETs in a rangefrom earth ground to Vsupply minus the voltage drop across therespective power MOSFET. The gate drive blocks 128 a and 128 c apply alevel shifted voltage signal to the gate terminal of the associatedpower MOSFET with a range of zero to +6 volts relative to the floatingsource terminal voltage. In this way, a 0 to 5 Volt ground-referencedsignal from the control block 146 is translated into a 0-6 Volt signal(buffered for high current) relative to the potential at the sourceterminals of the power MOSFETs 130 a and 130 c.

Each of the gate drive blocks also provide logic for controlling thecross conduction lockout of the power MOSFETs. If both an upper andlower power MOSFET, e.g., power MOSFETs 130 a and 130 b, are conductingat the same time, then “shoot through” currents will flow from the inputpower supply to ground which may possibly destroy these power MOSFETs.The gate drive blocks prevent this condition by simultaneously examiningthe value of the gate terminal voltages for both the upper and lowerpower MOSFETs. When one of the gate drive blocks (upper or lower)detects an “on” voltage at the gate terminal of the associated MOSFET,then the other gate drive block is locked out from also applying the onvoltage to its associated gate terminal.

The gate drive blocks 128 a and 128 c provide for initializing a pair ofbootstrap capacitors 150 a and 150 b during startup (initialenergization) of the present invention. Bootstrap capacitor 150 a isconnected between the OUT LEFT terminal and the BOOST LEFT terminal. Asdiscussed above, the OUT LEFT terminal is also connected to the sourceterminal of the power MOSFET 130 a and the gate drive block 128 a. Inthis way, the bootstrap capacitor 150 a is charged by the diode 126 awhen the lower power MOSFET 130 b is conducting and the upper powerMOSFET 130 a is non-conducting. Once charged, the bootstrap capacitor150 a will provide a stable turn on voltage to the gate terminal of theupper power MOSFET 130 a even as the potential at the source terminal ofthe upper MOSFET is rising to approximately the same potential asVsupply. Similarly, the bootstrap capacitor 150 b is connected betweenthe OUT RIGHT terminal and the BOOST RIGHT terminal to performsubstantially the same function. Also, the lamp 106 and the tank circuit108 are coupled between the OUT LEFT terminal and the OUT RIGHT terminalof the H-bridge output circuit.

During initialization, i.e., startup, of the IC 104, the lower powerMOSFETs 130 b and 130 d are switched on (conduction) by gate driveblocks 128 b and 128 d so that charge is simultaneously provided to thebootstrap capacitors 150 a and 150 b. When the H-Bridge output circuitbegins to oscillate and supply electrical power to the CCFL, thebootstrap capacitors 150 a and 150 b will sequentially partiallydischarge and recharge during the normal switching cycle of the powerMOSFETs. The diodes 126 a and 126 b automatically recharge theirassociated bootstrap capacitors 150 a and 150 b when their associatedpower MOSFETs 130 a and 130 c are turned off in the switching cycle. Inthis way, the bootstrap capacitors enable the gate drive blocks 128 aand 128 c to provide an adequate and stable turn on voltage to the gateterminals of the associated MOSFETs 130 a and 130 c.

To minimize the effect of surge requirements on the battery and “inrush” current to the lamp, both a soft on and soft off are implemented.The term “soft on” is used to describe a gradual increase in systempower to the normal level and “soft off” describes a gradual decrease insystem energy from the normal level. The amount of energy delivered tothe system is related to the output pulse width and can be adjusted bysumming current at the Comp pin. Additional current forced into the Comppin results in wider pulse widths, while current pulled from the Comppin results in narrower pulse widths. Specifically, near the end of aburst, current is pulled from the Comp pin in order to reduce the widthsof the output pulses. In one embodiment, when the Comp pin isapproximately 50 mV above ground and the pulse widths are near minimumthe burst is allowed to terminate. When the next burst is initiated theComp pin is near ground resulting in initially narrow pulses. A currentis then sourced to the Comp pin resulting in a gradual increase in pulsewidth until normal operation is reached.

Importantly, the energy of the power delivered to the load during thebeginning of startup is different from that during normal operation.Specifically, in one embodiment, the width of the first pulse is largerthan that of normal operation. In one embodiment, the initial energypulse is twice as long as the pulses during normal operation. This hasbeen found to increase the ability of the inverter to strike the CCFLunder different environmental and equipment conditions.

The mux block 134 switches between the drain terminals of the powerMOSFETs 130 b and 130 d, so that the current flowing through the powerMOSFETs may be determined by the control logic block 146. The current isdetermined by measuring the voltage across the power MOSFETs when theyare on, i.e., conducting. The measured voltage is directly related tothe amount of current flowing through the power MOSFET by its “on”resistance, which is a known value. Since the mux block 134 switchesbetween the drain terminals of the power MOSFET that is turned on, themux block output voltage is proportional to the current (Isw) flowingthrough the particular MOSFET that is turned on. The mux block 134 is apair of analog switches that commutate between the drain terminals ofthe lower power MOSFETs.

A peak current (Ipk) comparator 138 has an input coupled to the outputof the mux block 134 and another input coupled to a predeterminedvoltage, e.g., 200 mV that is derived from the Vref signal. An output ofthe peak current comparator 138 is coupled to the control logic block146 and an on-time timer 142. The peak current comparator 138 outputindicates to the control logic block 146 when a predetermined maximumcurrent level has been exceeded. If the lamp 106 is extinguished orbroken, the current flowing through the power MOSFETs will build to arelatively high value as the IC 104 tries to drive the requested amountof power or current into the relatively low loss tank circuitcomponents. Since a relatively high current flowing into the tankcircuit's capacitor may result in a dangerously high voltage at thesecondary of a step-up transformer, the control logic block 146 willturn off the power MOSFET when this condition is indicated by the peakcurrent comparator 138.

An input of a zero crossing detector 140 (comparator) is coupled to theoutput of the mux block 134 and another input is coupled to apredetermined voltage, e.g., 5 mV that is derived from the Vref signal.The output of the zero crossing detector 140 is coupled to the controllogic block 146 for indicating when the current in the tank circuit hasalmost fallen to zero Amps. The control logic block 146 uses the outputof the zero crossing detector 140 to determine when the rest phaseshould be terminated and initiate the next power phase in the cycle,e.g., power phase A or power phase B as presented in the discussion ofFIG. 6 below.

The on-time timer 142 determines the duration of each power phase forthe control logic block 146. One input to the on-time timer 142 iscoupled to an end of a loop compensation capacitor 148 and the output ofthe power control block 136. Another end of the loop compensationcapacitor 148 is coupled to the Vref signal. The on-time timer 142determines the period of time (duration) for each power phase inaccordance with the value of the voltage on the loop compensationcapacitor 148. The on-time timer 142 is separately coupled to an inputand an output of the control logic block 146 and the output of the peakcurrent (Ipk) comparator 138. Also, the on-time timer 142 will indicateto the control logic block 146 when the period of time for each powerphase has elapsed. In one embodiment, the on-time timer 142 is operativeto provide a pulse width 2.5 times as long as the pulses in normaloperation.

The brightness opamp 144 has an output coupled to a power control(analog multiplier) block 136. An input to the brightness opamp 144 iscoupled to a user selectable potentiometer (not shown) for receiving avoltage related to the setting of the potentiometer. When the userselects a control associated with the potentiometer a voltage isimpressed by the brightness opamp's output at the power control block136 that either proportionally increases or decreases in relation to thedisposition of the control. Further, as the voltage is changed by theuser selecting the control, the on-time timer 142 will indicate acorresponding change in the period of time for each power phase to thecontrol logic block 146.

The power control block 136 provides a signal as an input to a summingnode 141 that also inputs a reference current from a constant current(Iref) source 170. The output of the summing node 141 is coupled to theon-time timer 142 and an end of the loop compensation capacitor 148.

The switching of the mux block 134 is coordinated by the control logicblock 146, so that only one power MOSFET current at a time is measured.Also, the control logic block 146 measures the currents flowing throughthe lower H-bridge power MOSFETS 130 b and 130 d to synchronize thepower phase of the present invention with the current of the tankcircuit, determines when the current flowing through the power MOSFETshas exceeded a predetermined maximum peak current (Ipk), and computesthe actual power that is delivered to the load.

Generally, there are two types of cycle phases that the control logicblock 146 manages, i.e., the power phase and rest phase. The power phaseoccurs when diagonally opposed power MOSFETs are conducting. Forexample, power phase A occurs when the power MOSFETs 130 a and 130 d areon. Similarly, power phase B occurs when power the MOSFETs 130 b and 130c are on. In both power phases, the control logic block 146 will enablecurrent to flow through the power MOSFETs until one of the followingevents is indicated: (1) the peak current (Ipk) comparator 138 detectsthat the maximum current limit is exceeded such as when the dischargelamp is out; (2) the on-time timer 142 has timed out; or (3) the zerocrossing detector 140 provides an indication to the control logic block146 to switch the state of the MOSFETs to the next power phase in thecycle.

In a typical embodiment, when the on-time timer 142 has timed out inpower phase A, the control logic block 146 will switch the power MOSFETsto the rest phase. In the rest phase, the lower H-bridge power MOSFETs130 b and 130 d turn on and both upper H-bridge power MOSFETs 130 a and130 c turn off. Although the tank (output) circuit 108 coupled to theOUT LEFT and OUT RIGHT terminals may continue to provide current to theCCFL 106 for a brief period of time, the tank circuit's current willrapidly return to zero at which point the zero crossing detector 140will indicate this zero current condition to the control logic block146. Next, the control logic block 146 will direct power MOSFETs 130 cand 130 b to turn on and power MOSFETs 130 a and 130 d to turn off. Thecontrol logic block 146 continuously cycles the power MOSFETs from thepower phase A to the rest phase to the power phase B to the rest phaseand back to the power phase A at the resonant frequency of the load. Thecontrol logic block controls the amount of power/current driving thedischarge lamp by varying the amount of time spent resting (rest phase)in relation to the amount time spent adding energy (power phase) to thetank circuit.

Another embodiment provides for the control logic block 146 to use theindication from the peak current comparator 140 to determine when toswitch between phases. In this case, the control logic block 146 directsthe power MOSFETs to directly toggle (switch) between the A and B powerphases so that the rest phase is skipped entirely. In this mode ofoperation, the current waveform into the tank circuit has a triangularshape because the control logic block 146 actively drives the tankcircuit's current back the other way when the peak current comparator140 indicates that the “peak” current has been reached. This embodimentserves to constrain/control the current provided by the tank circuit 108and limit the open circuit voltage at the discharge lamp terminals.Either embodiment may be selected during the manufacture of the IC 104with a simple metal mask option.

There are at least two asynchronous digital logic inputs to the controllogic block 146 and they include: (1) a chip enable input for turningthe IC 104 on or off; and (2) a thermal shutdown input that provides forinternal thermal protection of the IC 104. Another digital input to thecontrol logic block 146 is a multifunctional test/burst input. Inproduct testing of the IC 104, this input is used to halt the executionof the start up initialization steps so that various parameters of theIC may be tested. However, once the product testing is complete, thisdigital logic input may be used to implement “burst mode” dimming.

In burst dimming mode, the user drives the burst input with arectangular logic waveform, in one state this input commands the IC 104to operate normally and deliver power to the lamp 106. In the otherstate the burst input causes the IC 104 to suspend normal operation andstop delivering power to the lamp 106. The burst input is normallyswitched off and on at a fast enough rate to be invisible (typically onthe order of 180 Hz or greater) for dimming the light emitted by thelamp 106. When the burst dimming mode is asserted, the loop compensationcapacitor 148 stops recharging or discharging, i.e, the voltageimpressed on the loop compensation capacitor 148 is saved so that theproper power level is quickly resumed when the burst dimming mode isde-asserted. Also, in the burst dimming mode, a relatively greater rangeof dimming for the lamp 106 is provided than a range provided by atypical analog dimming mechanism because the effect of parasiticcapacitances is reduced.

Additionally, full output and analog dimming is supported by the IC 104with other inputs to the control logic block 146 such as inputs from thepeak current (Ipk) comparator 138, the on-time timer 142, and the zerocrossing detector 140.

FIG. 4 illustrates an exemplary schematic 143 of the components employedto control the operation of the IC 104 with the amount of power drivingthe tank circuit 108. Since losses in the tank circuit 108 and thetransformer 114 are approximately constant over the entire range of theAC signal driving the load, the input power to the load correlates tothe actual power driving the CCFL 106 in the tank circuit 108. Also, thepower control block 136 is a metal mask option that must be selectedduring the manufacture of the IC 104.

Making use of the logarithmic relationship between the base-emittervoltage (Vbe) and collector current (Ic) of a bipolar transistor, asimple multiplier is implemented in the following manner. In one portionof the power control block 136, an end of a resistor 166 is coupled tothe Vsupply signal and another end is coupled to a drain terminal of aMOSFET 168. A gate terminal of the MOSFET 168 is coupled to the outputof the on-time timer 142 (not shown here). The on-time timer 142modulates the duty cycle of the current through the MOSFET 168 bycontrolling the voltage at the gate terminal synchronous with the outputpower phase waveform. A source terminal of the MOSFET 168 is coupled toa base of an NPN transistor 150, a base of an NPN transistor 156, and acollector of an NPN transistor 152. A collector of the NPN transistor150 is connected to the Vref signal. An emitter of the NPN transistor150 is coupled to a base of the NPN transistor 152 and a collector of anNPN transistor 154. An emitter of the NPN transistor 152 is coupled toground and an emitter of the NPN transistor 154 is coupled to an end ofa resistor 162 and an inverting input to an opamp 149. Another end ofresistor 162 is connected to ground. Also, a non-inverting input to theopamp 149 is coupled to the output from the mux block 134 (not shownhere) and an output of the opamp is coupled to a base of the NPNtransistor 154.

In another portion of the power control block 136, an emitter of the NPNtransistor 156 is coupled to a base of an NPN transistor 158 and acollector of an NPN transistor 160. An emitter of the NPN transistor 158is coupled to ground and a collector is coupled an end of the loopcompensation capacitor 148 and an output of a constant current (Iref)source 170. The other end of the loop compensation capacitor 148, aninput to the constant current (Iref) source 170 and a collector of theNPN transistor 156 are coupled to the Vref signal. An emitter of the NPNtransistor 160 is coupled to one end of a resistor 164 and the invertinginput to the brightness opamp 144. Another end of the resistor 164 isconnected to ground. A base of the NPN transistor 160 is coupled to anoutput of the brightness opamp 144. Although not shown, thenon-inverting input to the brightness opamp 144 is coupled to apotentiometer for enabling a user to “dim” the amount of light emittedby the lamp 106.

In the following analysis (description) of the operation of the powercontrol block 136, certain quantities may be neglected, compared toother, more significant quantities without compromising the results ofthe analysis. In particular, the various NPN transistor base currentsare neglected compared to the NPN transistor collector currents. Also,the supply voltage is assumed to be large compared to the sum of thebase-emitter voltages of the NPN transistor 150 and the NPN transistor152.

The power control block 136 determines the amount of power delivered tothe load by measuring a corresponding amount of power drawn from thepower supply. Also, the current either into or out of the loopcompensation capacitor 148 is the difference of a constant and amultiply and divide performed in the power control block 136.

During a power phase, the first multiplication is created when theon-time timer 142 supplies the turn on voltage to the gate terminal ofthe MOSFET 168 which causes the NPN transistors 150 and 152 to conductand provide a turn-on voltage to the base of the NPN transistor 156.Also, the opamp 149 will cause the NPN transistor 154 to conduct acurrent proportional to the output power switch current when the muxblock 134 has switched a drain terminal voltage (Vswitch) from theselected lower power MOSFET to the input of the opamp.

The collector current of the NPN transistor 150 is equal to thecollector current of the NPN transistor 154. Similarly, the collectorcurrent of the NPN transistor 152 is equal to the supply voltage(Vsupply) divided by the resistor 166. The base-emitter voltage of theNPN transistor 150 is proportional to the logarithm of the current inthe output switch. Similarly, the base-emitter voltage of the NPNtransistor 152 is proportional to the logarithm of the supply voltage.Thus, the voltage (with respect to ground) at the base terminal of theNPN transistor 150 is proportional to the logarithm of the product ofVsupply times Iswitch. It is important to note that this voltage ischopped, i.e., gated, by the duty cycle of the output waveform.

The voltage at the base of the NPN transistor 150 is equal to thevoltage at the base terminal of the NPN transistor 156. The collectorcurrent of the NPN transistor 160 is proportional to the(externally-provided) brightness control voltage. Also, the collectorcurrent of the NPN transistor 156 is equal to the collector current ofthe NPN transistor 160. Furthermore, the base-emitter voltage of the NPNtransistor 156 is proportional to the logarithm of the brightnesscontrol voltage. Thus, the voltage (with respect to ground) at the baseterminal of the NPN transistor 158 is proportional to the logarithm of(Vsupply*Iswitch/Vbright).

The collector current of the NPN transistor 158 must be proportional tothe anti-logarithm of its base voltage, i.e., the collector current ofthe NPN transistor 158 is proportional to (Vsupply*Iswitch/Vbright). Thecollector current of the NPN transistor 158 is averaged by the loopcompensation capacitor 148. The action of the control loop ensures thatthe average of the collector current of the NPN transistor 158 is equalto the constant current (Iref) source 170.

For example, when (Vsupply*Iswitch*duty cycle)>(Iref*Ibrt), extracurrent flows into the loop compensation capacitor 148 at the COMPterminal from the constant current (Iref) source 170, which has theeffect of shortening the duty cycle provided by the on-time timer 142and reducing the power supplied to the load. However, if(Vsupply*Iswitch*duty cycle)<(Iref*Ibrt), the loop compensationcapacitor 148 will discharge slightly and the on-time timer 142 willincrease the length of the duty cycle until the power drawn from theVsupply is equal to the power demanded by the control voltage at thenon-inverting input to the brightness amplifier. The integrated circuit104 modulates the duty cycle on the MOSFET 168 and the power MOSFETs 130a, 130 b, 130 c and 130 d until the voltage on the COMP terminal stopschanging. In this way, negative feedback at the COMP terminal is used tomodulate the duty cycle provided by the on-time timer 142.

FIG. 5 shows how, in addition to buffering the low current logicsignals, an exemplary gate drive block 128 b may also provide a localcurrent limit for the associated power MOSFET 130 b while it is on. Aninput to the gate drive block 128 b is coupled to an input of a one shottimer 170, a reset input to an R-S flip-flop 172 and an input to an ANDgate 174. An output of the flip-flop 172 is coupled to another input tothe AND gate 174 and the set input of the flip-flop is coupled to anoutput of an AND gate 176. The output of AND gate 174 is coupled to aninput of an inverter 178 that has an output connected to the gate of theMOSFET 130 b. An output of the one shot timer 170 is connected to aninput to the AND gate 176. A current limit comparator 180 has an outputconnected to another input to the AND gate 176. One input to thecomparator 180 is coupled to an approximately 50 millivolt signalderived from the Vref signal and another input is coupled to the sourceterminal of the MOSFET 130 b and an end of a resistor 182. The value ofthe resistor 182 is sized to provide a predetermined voltage at theinput to the comparator 180 when five or more Amps of current areflowing through the resistor to ground.

The one shot timer 170 provides a signal approximately 200 nanosecondsafter the power MOSFET 130 b has turned on during the power phase (longenough for the switching noise to stop). The output signal of the oneshot timer 170 enables the output of the current limit comparator 180 tobe provided by the AND gate 176 to the set input of the flip-flop 172.If the output of the current limit comparator 180 indicates that thecurrent limit voltage on the resistor 182 has been reached, theflip-flop will output a turn-off signal to the AND gate 174 which inturn outputs the turn-off signal to the inverter 178 so that a turn offvoltage is applied to the gate terminal of the MOSFET 130 b. In thisway, the power MOSFET 130 b is immediately turned off for the remainderof a power phase when a current greater than five Amps flows through thepower MOSFET. Similarly, the gate drive block 128 d provides forlimiting the current flow through the MOSFET 130 d in substantially thesame way.

FIG. 3B shows an exemplary schematic of a current control embodiment ofthe invention as implemented by an IC 104′. Although the schematic ofthe current control IC 104′ is similar to the power control IC 104,there are some differences. Since current control is employed by the IC104′ to regulate the electrical power supplied to the lamp 106, thepower control block 136 is not provided in the IC 104′. Also, the outputof the brightness opamp 144 is provided to the summing node 141 whichalso receives an Isense current through a connection to the senseresistor 109 as shown in FIG. 1B. Similarly, the output of the summingnode 141 is provided to the end of the loop compensation capacitor 148and the on-time timer 142. The current through the sense resistor 109proportionally approximates the amount of current flowing through thelamp 106. The IC 104′ uses this approximation to control the amount ofelectrical power driving the lamp 106.

The current control version of the IC 104′ uses the brightness opamp 144to convert the user input at the potentiometer into a current (Ibright)that the summing node 141 compares to the Isense current and the currentdifference flows either into or out of the loop compensation capacitor148. In contrast, the power control version of the IC 104 performs thefollowing generalized steps: (1) employ the brightness opamp 144 toconvert the user input into the Ibright current; (2) use the analogmultiplier to logarithmically add (multiply) currents proportional tothe Iswitch current, Vsupply and the duty cycle; (3) employ the analogmultiplier to logarithmically subtract (divide) the Ibright current fromthe logarithmically added currents; (4) compare the result of theantilogarithm of the subtraction to the Ireference current to determinea differential current; and (5) employ the differential current toeither charge or discharge the compensation capacitor 148 so that theon-time timer 142 will adjust the time interval of each power phaserelative to the voltage impressed across the loop compensation capacitor148 by the amount of stored charge.

Looking now to FIG. 6, a schematic overview 200 shows the presentinvention configured in four operational modes or phases that complete acycle for driving a load with an AC signal. All four phases, i.e., apower phase “A” 202, a rest phase “A” 204, a power phase “B” 206, and arest phase “B” 208, employ the same components. The power MOSFETs 130 a,130 b, 130 c and 130 d are illustrated as discrete switches. When apower MOSFET is on (conducting), it is represented as a closed switch.Also, when the power MOSFET is off (non-conducting), it is representedas an open switch. In this way, the state of conduction for the powerMOSFETs may be more clearly illustrated for the different phases of thecycle.

An end of the power transistor 130 a is connected to the Vsupplyterminal and the other end is coupled to an end of the power transistor130 b and an end of the tank circuit 108. One end of the powertransistor 130 c is connected to the Vsupply signal (DC supply) and theother end is connected to the other end of the tank circuit 108 and oneend of the power transistor 130 d. The other ends of power transistors130 b and 130 d are connected to ground.

As illustrated in power phase “A” 202, diagonally opposite powertransistors 130 b and 130 c are off (open position) and powertransistors 130 a and 130 d are on (closed position). A DC current fromthe Vsupply terminal flows through the power transistor 130 a, passesthrough the tank circuit 108 and returns to earth ground through powertransistor 130 d.

When the flow of current from the Vsupply terminal is at least equal toa predetermined peak value as indicated by the peak current comparator138 or the on-time timer 142 has finished, the power transistors willswitch from power phase “A” 202 to the configuration identified as restphase “A” 204. However, if neither of these conditions has occurred andthe tank current has returned to the zero crossing point as indicated bythe zero crossing detector 140, the power transistors will bypass therest phase “A” and switch directly to the configuration identified as apower phase “B” 206. Typically, the bypassing of the rest phase willoccur when there is a high load and a relatively low Vsupply voltage.

Rest phase “A” 204 is shown with the top laterally opposite powertransistors 130 a and 130 c disposed in the open position (off) and thebottom laterally opposite power transistors 130 b and 130 d configuredin the closed position (on). In the rest phase “A” 204 configuration,the tank circuit 108 discharges stored energy, i.e., a current, throughpower transistor 130 d to ground. After the tank circuit has dischargedat least a portion of its stored energy, the power transistors switch tothe configuration identified as the power phase “B” 206. The presentinvention provides for tracking the resonant frequency of the tankcircuit and switching the power transistors at this frequency, so thatthe tank circuit will store energy during the power phase “A” 202 anddischarge this energy during the rest phase “A”. In this way, the ACsignal impressed across the load coupled to the tank circuit has arelatively smooth sinusoidal shape for the “A” portion of the AC signalcycle.

Similarly, the power phase “B” 206 illustrates diagonally opposite powertransistors 130 a and 130 d disposed in an open position and powertransistors 130 b and 130 c in a closed position. A current from theVsupply terminal flows through the power transistor 130 c, passesthrough the tank circuit 108 and returns to ground through powertransistor 130 b. When the flow of current from the Vsupply terminal isat least equal to a predetermined peak current value indicated by thepeak current comparator 138 or the on-time timer 142 has timed out, thepower transistors switch from the power phase “B” 206 to theconfiguration identified as the rest phase “B“208.

Rest phase “B” 208 is shown with top laterally opposite powertransistors 130 a and 130 c disposed in the open position and the powertransistors 130 b and 130 d configured in the closed position. In therest phase “B” 208, the tank circuit 108 will discharge stored energy,i.e., a current, through power transistor 130 b to ground so that the ACsignal impressed across the load coupled to the tank circuit has asmooth sinusoidal shape for the “B” portion of the AC signal cycle.After discharging the stored energy for a period of time, the powertransistors will switch to the power phase “A” configuration and thecycle of phases will repeat. In this way, power is transferred to theload continuously throughout the cycle (both power and rest phases) andthe stored energy in the tank circuit 108 is replenished during eachpower phase.

The present invention provides for dimming a lamp, i.e., reducing theamount of power delivered to the load, by decreasing the period of timethat the power transistors are disposed in the power phase “A” and thepower phase “B” configuration and proportionally increasing the periodof time that the transistors are disposed in the rest phase “A” and therest phase “B” positions.

Under normal operating conditions, the lamp current (or power) ismeasured and compared in a feedback loop to the user input (setting ofthe potentiometer). An error (difference) between the measured value ofthe lamp current and the user input is employed to determine the valueof the voltage across the loop compensation capacitor 148 that issubsequently employed by the on-time timer 142 to determine the lengthof time that the power transistors are turned on for the power phases.In this way, the user may control the brightness of the lamp 106 over arelatively large range by adjusting the setting of the potentiometer.

FIG. 7A-7D includes four graphs that illustrate the correspondencebetween a AC voltage signal generated by the present invention and thecurrent supplied to the load, i.e., the CCFL, under maximum power andreduced power conditions. In a top row graph 210, a horizontal time axis216 and a vertical voltage axis 218 are shown. As is typical of anH-bridge configuration, the peak voltage amplitude 212 and 214 is equalto the voltage provided by the power supply and the peak-to-peak loadvoltage is twice the supply voltage. A substantially straight, verticalrising edge 220 occurs at the zero crossing of the tank circuit'scurrent each time the negative waveform 214 transitions to the positivewaveform 212. Similarly, a vertical falling edge 222 occurs when thepower phase terminates for one of the three reasons that power phasesterminate as discussed above. Additionally, the graph 210 shows thevoltage waveform shape when the IC 104 is delivering the maximumpower/current to the tank circuit for each of the half cycles of thetank's resonant frequency. Typically, this waveform is observed when thecircuit is delivering design maximum power to the load at design minimumsupply voltage.

In a second row graph 230 of FIG. 7B, a horizontal time axis 232 and avertical current axis 224 are displayed that correspond to the voltagewaveform illustrated in the graph 210. The maximum value of the positivecurrent waveform 226 is equal to a positive peak current value.Similarly, the maximum value of the negative current waveform 228 isequal to a negative peak current value. A rounded falling edge 234occurs at the resonant frequency of the tank circuit 108 when thepositive current waveform 226 has finished charging up the circuit.Similarly, a rounded rising edge 235 occurs at the resonant frequency ofthe tank circuit 108 when the circuit is just beginning to charge up.

In a third row graph 240 of FIG. 7C, a horizontal time axis 242 and avertical voltage axis 244 are displayed. The peak voltage amplitudedelivered to the load by the voltage waveforms 236 and 238 are equal tothe supply voltage and the peak-to-peak load voltage is twice the supplyvoltage. In graph 240, the duty cycles of both the positive-goingwaveforms 236 and the negative-going waveforms 238 have been reduced toabout one third of the maximum duty cycle (100%). The graph 240illustrates trailing-edge modulation of the duty cycle of the drivingwaveform, i.e., the leading edge of the voltage pulse of both polaritiesoccurs near the zero-crossing of the current waveform for all values ofthe duty cycle. Also, graph 240 shows the case of the voltage providedby the power supply not delivering the maximum power capacity of theH-bridge circuit such as when the lamp is dimmed or the power supplyvoltage is higher than the design minimum value. In contrast, the graph210 shows the case of the maximum amount of delivered power matching themaximum capacity of the tank circuit.

In a fourth row graph 246 of FIG. 7D, a horizontal time axis 248 and avertical current axis 250 are displayed that correspond to the voltagewaveform illustrated in the graph 240. The maximum value of a positivecurrent waveform 252 is equal to the positive peak current value.Similarly, the maximum value of a negative current waveform 254 is equalto the negative peak current value. A rounded rising edge 256 occurs atthe resonant frequency of the tank circuit 108 when the positive currentwaveform 252 is charging up the circuit and when the circuit initiallybegins to discharge current to the load. Similarly, a rounded fallingedge 258 occurs when the tank circuit 108 starts to discharge lesscurrent to the load. It is important to note that the tank circuitprovides for smoothing the current waveform provided to the load whenthe voltage waveform is operating at less than a 100% duty cycle. Thevoltage waveform pulses shown in graph 240 pulse at the zero crossingpoint of the current waveform illustrated in the graph 246 so that theamount of energy delivered to the tank is controlled.

FIGS. 7E and 7F are two graphs that illustrate the correspondencebetween a leading edge modulation of the AC voltage signal generated bythe present invention and the current supplied to the load, underreduced power conditions. The leading edge modulation of the AC voltagesignal may be used in substantially the same manner as indicated inFIGS. 7A-7D for the trailing edge AC voltage signal. For leading edgemodulation, the AC voltage signal is turned on sometime after the zerocrossing point of the AC current waveform has occurred and turns off atits next zero crossing point.

In a top row graph 241 of FIG. 7E, a horizontal time axis 247 and avertical voltage axis 245 are shown. The peak voltage amplitudedelivered to the load by the voltage waveforms 237 and 239 are equal tothe supply voltage and the peak-to-peak load voltage is twice the supplyvoltage. In graph 241, the duty cycles of both the positive-goingwaveforms 237 and the negative-going waveforms 239 have been reduced toabout one third of the maximum duty cycle (100%). Also, graph 241 showsthe case of the voltage provided by the power supply not delivering themaximum power capacity of the H-bridge circuit such as when the lamp isdimmed or the power supply voltage is higher than the design minimumvalue.

In a bottom row graph 247 of FIG. 7F, a horizontal time axis 249 and avertical current axis 251 are displayed that correspond to the voltagewaveform illustrated in the graph 241. The maximum value of a positivecurrent waveform 253 is equal to the positive peak current value.Similarly, the maximum value of a negative current waveform 255 is equalto the negative peak current value. A rounded rising edge 257 occurs atthe resonant frequency of the tank circuit 108 when the positive currentwaveform 253 is charging up the circuit and when the circuit initiallybegins to discharge current to the load. Similarly, a rounded fallingedge 259 occurs when the tank circuit 108 starts to discharge lesscurrent to the load. The voltage waveform pulses shown in graph 241pulse before the zero crossing point of the current waveform illustratedin the graph 247 so that the amount of energy delivered to the tank iscontrolled.

In FIG. 8A, a graph 260 illustrates the double-sided phase modulation ofthe AC voltage signal. A vertical voltage (Vab) axis 264 and ahorizontal time axis 262 are displayed that correspond to the voltagewaveform illustrated in graph 260. In the H-bridge, the peak voltagepositive and negative waveforms 266 and 268 are equal to the supplyvoltage and the peak-to-peak voltage is twice the supply voltage. In asecond graph 271 of FIG. 8B, a horizontal time axis 267 and a verticalcurrent axis 265 are displayed which correspond to the voltage waveformillustrated in graph 260. The maximum value of a positive currentwaveform 270 is equal to the positive peak current value. Similarly, themaximum value of a negative current waveform 269 is equal to thenegative peak current value. Additionally, since double-sided phasemodulation centers the voltage waveform at the peak of the correspondingcurrent waveform, the present invention provides for either increasingor decreasing the width (both sides) of the voltage waveform in relationto the amount of power delivered to the load.

In FIG. 9A-9D, four graphs illustrate pulse train phase modulation ofthe AC voltage signal and the current supplied to the load under maximumpower conditions. In a top row graph 278 of FIG. 9A, a horizontal timeaxis 272 and a vertical voltage axis 274 are shown. A positive voltagesquare-shaped waveform 276 is equal to the voltage provided by thevoltage supply. Also, the waveform is on for the first half of the powercycle and off for the second half of the cycle.

In a second row graph 286 of FIG. 9B, a horizontal time axis 284 and avertical voltage axis 280 are shown. A positive voltage square-shapedwaveform 282 is equal to the voltage provided by the voltage supply.Also, the waveform is off for the first half of the power cycle and onfor the second half of the cycle.

In a third row graph 288 of FIG. 9C, a horizontal time axis 296 and avertical voltage axis 290 are shown. A positive voltage square-shapedwaveform 292 is equal to the voltage provided by the voltage supply anda negative voltage square-shaped waveform 294 is equal to the voltageprovided by the supply. Also, the voltage waveforms alternate being onduring the power cycle, i.e., the positive waveform is on for the firsthalf of the cycle and the negative waveform is on for the second half.

In a fourth row graph 300 of FIG. 9D, a horizontal time axis 302 and avertical current axis 306 are displayed that correspond to the voltagewaveform illustrated in the graph 288. The maximum value of the positivecurrent waveform 304 is equal to a positive peak current value.Similarly, the maximum value of the negative current waveform 303 isequal to a negative peak current value.

In FIGS. 9E-9H, four graphs illustrate pulse train phase modulation ofthe AC voltage signal and the current supplied to the load under reducedpower conditions. In a top row graph 308 of FIG. 9E, a horizontal timeaxis 310 and a vertical voltage axis 312 are shown. A positive voltagesquare-shaped waveform 314 is equal to the voltage provided by thevoltage supply. Also, the positive waveform 314 has a 50 percent dutycycle, i.e., the waveform is on for the first and second quarters (firsthalf) of the power cycle and off for the third and fourth quarters(second half) of the cycle.

In a second row graph 318 of FIG. 9F, a horizontal time axis 320 and avertical voltage axis 322 are shown. A positive voltage square-shapedwaveform 316 is equal to the voltage provided by the voltage supply.Also, the positive voltage waveform has a 50 percent duty cycle, i.e.,the waveform is on for the second and third quarters of the power cycleand off for the first and fourth quarters of the cycle.

In a third row graph 326 of FIG. 9G, a horizontal time axis 328 and avertical voltage axis 324 are shown. A positive voltage square-shapedwaveform 330 is equal to the voltage provided by the voltage supply anda negative voltage square-shaped waveform 333 is equal to the voltageprovided by the supply. The positive voltage waveform 330 is only on forthe first quarter of the power cycle and the negative waveform 333 isonly on for the third quarter of the cycle. During the second and fourthquarters of the power cycle, the net voltage across the load is zerobecause the voltage at the two outputs of the H bridge are equal andtherefore cancel each other out.

In a fourth row graph 336 of FIG. 9H, a horizontal time axis 338 and avertical current axis 340 are displayed that correspond to the voltagewaveform illustrated in the graph 326. The maximum value of the positivecurrent waveform 342 is equal to a positive peak current value.Similarly, the maximum value of the negative current waveform 343 isequal to a negative peak current value. Also, the current waveform isshown delivering a reduced amount of power to the load. Additionally, itis envisioned that the relative phase of the voltage waveforms shown ingraphs 308 and 318 could be varied to further modulate the amount ofpower delivered to the load.

Looking now to FIG. 10, a schematic overview 344 shows the presentinvention configured in four operational modes that complete a cycle fordriving a load with a phase modulated AC signal. All four phases, i.e.,a power phase “I” 346, a rest phase “II” 348, a power phase “III” 350,and a rest phase “IV” 352, employ the same components. The power MOSFETs130 a, 130 b, 130 c and 130 d are illustrated as discrete switches. Whena power MOSFET is on (conducting), it is represented as a closed switch.Also, when the power MOSFET is off (non-conducting), it is representedas an open switch. In this way, the state of conduction for the powerMOSFETs may be more clearly illustrated for the different phases of thecycle. The physical configuration of the MOSFETs is substantiallysimilar to the configuration as presented in the discussion of FIG. 10above.

As illustrated in power phase “I” 346, diagonally opposite powertransistors 130 b and 130 c are off (open position) and powertransistors 130 a and 130 d are on (closed position). A current from theVsupply terminal flows through the power transistor 130 a, passesthrough the tank circuit 108 and returns to earth ground through powertransistor 130 d.

When the flow of current from the Vsupply terminal is at least equal toa predetermined peak value as indicated by the peak current comparator138 or the on-time timer 142 has finished, the power transistors willswitch from power phase “I” 346 to the configuration identified as restphase “II” 348. However, if neither of these conditions has occurred andthe tank current has returned to the zero crossing point as indicated bythe zero crossing detector 140, the power transistors will bypass therest phase “A” and switch directly to the configuration identified as apower phase “III” 350. Typically, the bypassing of a rest phase willoccur when there is a high load and a relatively low Vsupply voltage.

Rest phase “II” 348 is shown with the top laterally opposite powertransistors 130 a and 130 c disposed in the closed position (on) and thebottom laterally opposite power transistors 130 b and 130 d configuredin the open position (off). In the rest phase “II” 348 configuration,the tank circuit 108 discharges stored energy into the load bycirculating a current through power transistors 130 a and 130 c. Afterthe tank circuit has discharged at least a portion of its stored energy,the power transistors switch to the configuration identified as thepower phase “III” 350.

Similarly, the power phase “III” 350 illustrates diagonally oppositepower transistors 130 a and 130 d disposed in an open position and powertransistors 130 b and 130 c in a closed position. A current from theVsupply terminal flows through the power transistor 130 c, passesthrough the tank circuit 108 and returns to ground through powertransistor 130 b. When the flow of current from the Vsupply terminal isat least equal to a predetermined peak current value indicated by thepeak current comparator 138 or the on-time timer 142 has timed out, thepower transistors switch from the power phase “III” 350 to theconfiguration identified as the rest phase “IV” 352.

Rest phase “IV” 352 is shown with top laterally opposite powertransistors 130 a and 130 c disposed in the open position and the powertransistors 130 b and 130 d configured in the closed position. In therest phase “B” 208, the tank circuit 108 will discharge stored energy,i.e., a current, through power transistor 130 b to ground. Afterdischarging the stored energy for a period of time, the powertransistors will return to the power phase “I” 346 configuration and thecycle of phases will repeat. In this way, power is transferred to theload continuously throughout the cycle (both power and rest phases) andthe stored energy in the tank circuit 108 is replenished during eachpower phase.

In burst mode dimming, the discharge lamp 106 is switched on and off atan invisibly fast rate such as 180 Hertz. When the discharge lamp 106 ison, the frequency of the AC signal driving the lamp is determined by theon-time timer 142 and the zero crossing detector 140. A typicaloperating frequency would be 50 kilohertz. For a 50% burst mode dimming,the discharge lamp 106 would be turned off half of the time. In practicefor the representative frequencies chosen this would mean that an ontime would last 2.7 milliseconds and would comprise 135 cycles of 50 khzoscillation. This on time would be followed by 2.7 milliseconds of offtime. Similarly, a 5% burst mode dimming would have an on time of 0.27milliseconds comprising about 13 cycles of 50 Khz lamp current followedby approximately 5.3 milliseconds of off time. The sum of the on and offperiods would equal 180 hertz. When burst mode dimming is asserted (thedischarge lamp is off), analog feedback in the IC 104 is consideredinvalid. In this way, the loop compensation capacitor 148 is neithercharged nor discharged and the correct on-time setting for the on-timetimer 142 is “remembered” between burst mode off states.

The foregoing provides a detailed description of one particularembodiment of the present invention. However, in the more general sense,the method of the present invention is shown in FIG. 11. First, at step1101, the inverter 100 is initialized. This may, for example, includevarious procedures such as powering up various components. Next, at step1103, the inverter 100 provides one or more high energy pulses in orderto strike the lamp (or other load). The term “high energy pulses” asused herein refers to a pulse of energy that is higher than that of theenergy pulses during normal operation. In the pulse width modulationdescribed above, this corresponds to a wider pulse width. Finally, atstep 1105, after the lamp has been struck, the high energy pulses arediscontinued and normal energy pulses are provided.

While the preferred embodiment of the invention has been illustrated anddescribed, it will be appreciated that various changes can be madetherein without departing from the spirit and scope of the invention.

1-26. (canceled)
 27. A method of controlling a DC to AC inverter thatincludes a network of plurality of switches that generates an AC signalfrom a DC signal, a tank circuit, and a controller for driving adischarge lamp, comprising: initializing said controller; striking saiddischarge lamp using a first AC signal generated by said network ofplurality of switches and said tank circuit, said first AC signal havingthe same frequency as a resonant frequency of said tank circuit; anddriving said discharge lamp using a second AC signal generated by saidnetwork of plurality of switches and said tank circuit, wherein saidfirst AC signal has a higher electrical energy than said second ACsignal.
 28. The method of claim 1 wherein said first AC signal has alarger pulse width than said second AC signal.
 29. The method of claim 2wherein said driving said striking said discharge lamp using a first ACsignal further comprises modulating the pulse width of each half cycleof said first AC signal.
 30. The method of claim 1 wherein said first ACsignal has a larger voltage amplitude and the same pulse width with saidsecond AC signal.
 31. The method of claim 1 further comprising:monitoring an electrical power delivered to said discharge lamp using afeedback control mechanism; and adjusting said electrical powerdelivered into said discharge lamp whenever said electrical energy isoutside of a predetermined range.
 32. The method of claim 1 wherein saiddriving said discharge lamp further comprises: monitoring an on and offstate of each of said switches in said network of plurality of switches;and controlling said on and off state of said network of switches toavoid a shoot-through problem.
 33. The method of claim 1 wherein saiddriving said discharge lamp further comprising using parasiticcapacitances and inductances to generate said resonant frequency in saidtank circuit.
 34. The method of claim 1 wherein said driving saiddischarge lamp further comprises: determining whether a thermal overloadcondition occurs at said discharge lamp; and terminating said second ACsignal that drives said discharge lamp.
 35. The method of claim 1wherein said driving said discharge lamp further comprises: detecting anopen-circuit condition in said discharge lamp; and reducing the powerdelivered to said discharge lamp after said open-circuit condition. 36.The method of claim 1 wherein said driving said discharge lamp furthercomprises: detecting whether an electrical malfunction condition existsin said discharge lamp; and terminating said second AC signal deliveredto said discharge lamp after said electrical malfunction is detected.37. An apparatus for driving a discharge lamp, comprising: means forconverting a DC signal into an AC signal; means for filtering said ACsignal and for oscillating said AC signal so that said AC signal has afrequency based on a resonant frequency of said filtering means; andmeans for controlling said converting means and for providing a higherelectrical energy to said discharge lamp during striking of saiddischarge lamp relative to a normal operation, said filtering meanselectrically coupled between said converting means and said controllingmeans.
 38. The apparatus of claim 11 further comprises means formonitoring an electrical energy in said lamp and for feeding saidelectrical energy to said controlling means so that said controllingmeans adjusts the amount of said electrical energy provided to saiddischarge lamp.
 39. The apparatus of claim 11 wherein said controllingmeans further comprising means for monitoring said converting means,said monitoring means electrically coupled to control said convertingmeans so as to avoid shoot-through problems in said converting means.